Planar filter, semiconductor device and radio unit

ABSTRACT

A planar filter has first and second U-shaped open transmission line resonators ( 103, 105 ) and a crank-shaped open transmission line resonator ( 104 ), so that it is possible to decrease an area to be virtually occupied by the filter on a dielectric substrate ( 110 ) and enhance the attenuation characteristic.

This application is the US national phase of International ApplicationPCT/JP2003/014617 filed Nov. 18, 2003, which designated the US.PCT/JP2003/014617 claims priority to JP Patent Application No.2002-340506 filed Nov. 25, 2002 and JP Patent Application No.2003-359073 filed Oct. 20, 2003. The entire contents of theseapplications are incorporated therein by reference.

BACKGROUND OF THE INVENTION

The present invention relates to a planar filter suitable for use, forexample, in microwave bands including millimeter wave bands, and moreparticularly to a planar filter preferable for use in high-frequencyradio communication devices such as millimeter wave communicationdevices using a frequency of 30 GHz or more, as well as to asemiconductor device and a radio unit having the planar filter.

Conventionally, there have been planar filters that use microstripresonators. A design method thereof is described in, e.g., a literature“Basics and Applications of Microwave Circuits” by Yoshihiro Konishi,pages 369-373, published by Sogo Denshi Publishing, Aug. 20, 1990).

FIGS. 6A and 6B show one example of conventional planar filters. FIG. 6Ais a plan view and FIG. 6B is a cross sectional view taken along theline D-D′ in FIG. 6A. The planar filter is structured such that an inputline 1, an output line 2, a resonator 3, a resonator 4 and a resonator 5are formed on a dielectric substrate 10 having a grounding conductor 11on the back face. Each of the resonator 3, the resonator 4 and theresonator 5 has a line length that is half an equivalent wavelength of apassband center frequency.

As shown in FIG. 6A, a part of the input line 1 and a part of theresonator 3 are in parallel proximity to each other with a constant gaptherebetween to thereby establish electromagnetic coupling. Also, a partof the resonator 3 and a part of the resonator 4 are in parallelproximity to each other with a constant gap therebetween so as toachieve electromagnetic coupling. In a similar manner, the resonator 4and the resonator 5, as well as the resonator 5 and the output line 2are in parallel proximity to each other with a constant gap therebetweenfor electromagnetic coupling, respectively. By appropriately disposingthe resonators 3 to 5 and the input/output transmission lines 1, 2 tooptimize the degree of coupling, a desired bandwidth can be achieved.The shown planar filter has thee resonators 3, 4, and 5. It is to benoted that while a larger number of resonators can increase attenuationoutside the band, it also increases loss in the pass band and the areato be occupied by the filter.

The shape and arrangement of the resonators in the conventional planarfilter shown in FIG. 6 has following problems. If the resonators arearrayed in a longitudinal direction, the size of the planar filter isincreased. Particularly in the case where the planar filter isintegrated on an IC chip for reducing a loss in a connection sectionbetween the planar filter and other high-frequency integrated circuits,the conventional resonator layout deteriorates space efficiency of theIC chip and increases dead space not available for other circuits, withthe result that the size of the IC chip and the unit cost of the chipare increased.

SUMMARY OF THE INVENTION

In consideration of these drawbacks, an object of the present inventionis to provide a planar filter occupying a small area, suitable forintegration on an IC chip, and having good wave filtrationcharacteristics and good attenuation characteristics.

In order to accomplish the object, a planar filter according to thepresent invention has a first U-shaped open transmission line resonator,a second U-shaped open transmission line resonator, and a crank-shapedopen transmission line resonator.

In this invention, the provision of the first and second U-shaped opentransmission line resonators and the crank-shaped open transmission lineresonator makes it possible to decrease the area to be virtuallyoccupied by the filter on a dielectric and enhance the attenuationcharacteristic. This allows a device having the planar filter to bedownsized.

In one embodiment, the first and second U-shaped open transmission lineresonators and the crank-shaped open transmission line resonator have aline length that is half an equivalent wavelength of a passband centerfrequency component. This allows the planar filter to have an enhancedfiltration characteristic.

In one embodiment, the first and second U-shaped open transmission lineresonators and the crank-shaped open transmission line resonator arearranged so as to be electromagnetically coupled in an order of thefirst U-shaped open transmission line resonator, the crank-shaped opentransmission line resonator, and the second U-shaped open transmissionline resonator. The planar filter further comprises a first input/outputtransmission line and a second input/output transmission line, and thefirst input/output transmission line is arranged so as to beelectromagnetically coupled to the first U-shaped open transmission lineresonator, and the second input/output transmission line is arranged soas to be electromagnetically coupled to the second U-shaped opentransmission line resonator.

In this embodiment, due to the shapes and arrangement, or layout, of thefirst and second U-shaped open transmission line resonators and thecrank-shaped open transmission line resonator, it is possible todecrease the area to be virtually occupied by the filter on adielectric. This allows downsizing of a device having the planar filter.

In one embodiment, the first and second input/output transmission linesand the crank-shaped open transmission line resonator are arranged suchthat a part of at least one of the first and second input/outputtransmission lines and a part of the crank-shaped open transmission lineresonator are electromagnetically coupled to each other.

In this embodiment, part of the first and/or second input/outputtransmission line serving as an input transmission line or an outputtransmission line bypasses the first and second U-shaped opentransmission line resonators and establishes direct electromagneticcoupling with the crank-shaped open transmission line resonator.Consequently, in addition to a first transmission route on which asignal is transmitted in the order of the first input/outputtransmission line (input line), the first U-shaped open transmissionline resonator, the crank-shaped open transmission line resonator, thesecond U-shaped open transmission line resonator and the secondinput/output transmission line (output line), there is formed a secondtransmission route on which a signal is transmitted in the order of thefirst input/output transmission line (input line), the crank-shaped opentransmission line resonator, and the second input/output transmissionline (output line).

Therefore, appropriate adjustment of a phase difference between thefirst and the second transmission routes allows mutual cancellation ofsignals at frequencies in the close vicinity of the passband. Thisallows the attenuation characteristic outside the passband to be steep.

In one embodiment, the first and second U-shaped open transmission lineresonators and the crank-shaped open transmission line resonator areformed on a semiconductor substrate. This embodiment facilitatesfabrication of a semiconductor device having a small-sized,high-performance planar filter.

A semiconductor device according to an embodiment has theabove-described planar filter, which is integrated with a mixer on asemiconductor substrate. In this embodiment, the planar filter is formedon a semiconductor substrate in the integrated manner with the mixer, sothat power loss in a connection section between the mixer and the planarfilter can be minimized, which in turn allows a more compactsemiconductor device with higher performance to be realized.

Further, a radio unit in one embodiment has the planar filter. Since theradio unit in the embodiment has the planar filter, it becomes possibleto realize a radio communication device and a radio relay device as acompact and high-performance radio unit.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1A is a plan view showing a first embodiment of the planar filterof the present invention, FIG. 1B is a cross sectional view taken alongline A-A′ in FIG. 1A, and FIGS. 1C, 1D and 1E are views respectivelyshowing a first U-shaped open transmission line resonator, acrank-shaped open transmission line resonator and a second U-shaped opentransmission line resonator that the first embodiment has;

FIG. 2A is a plan view showing a second embodiment of the planar filterof the present invention, FIG. 2B is a cross sectional view taken alongline A-A′ in FIG. 2A, and FIGS. 2C, 2D and 2E are views respectivelyshowing a first U-shaped open transmission line resonator, acrank-shaped open transmission line resonator and a second U-shaped opentransmission line resonator that the second embodiment has;

FIG. 3 is a graph showing a frequency characteristic of the planarfilter in the second embodiment;

FIG. 4A is a plan view showing a planar filter-integrated even-harmonicmixer as a third embodiment of the present invention, and FIG. 4B is across sectional view taken along line C-C′ in FIG. 4A;

FIG. 5 is a block diagram showing a configuration example of a radiorelay device as a fourth embodiment employing the planar filter of thepresent invention;

FIG. 6A is a plan view showing an example of a conventional planarfilter, and FIG. 6B is a cross sectional view taken along line D-D′ inFIG. 6A;

FIG. 7 is a graph showing changes in a passing characteristic of theplanar filter in the second embodiment of the present invention in thecase where a gap between an input/output transmission line and thecrank-shaped open transmission line resonator is changed, to show theeffect of electromagnetic coupling between the input/output transmissionline and the crank-shaped open transmission line resonator;

FIG. 8 is a diagram showing IF signal frequency dependence of conversiongain of a desired wave and an undesired wave in a planarfilter-integrated even-harmonic mixer according to a third embodiment ofthe present invention; and

FIG. 9 is a block view showing a configuration example of a radio relaydevice as a fifth embodiment including the planar filter of the presentinvention.

DETAILED DESCRIPTION OF THE INVENTION

Hereinbelow, the present invention will be described in conjunction withthe embodiments with reference to the drawings.

First Embodiment

FIGS. 1A and 1B show a planar filter in the first embodiment of thepresent invention. FIG. 1A is a plan view, and FIG. 1B is a crosssectional view taken along line A-A′ in FIG. 1A. As shown in FIG. 1A,the planar filter in the first embodiment has a first input/outputtransmission line 101 serving as an input line, a second input/outputtransmission line 102 serving as an output line, a first U-shaped opentransmission line resonator 103, a second U-shaped open transmissionline resonator 105 and a crank-shaped open transmission line resonator104, which are formed on a dielectric substrate 110. As shown in FIG.1B, the dielectric substrate 110 has a grounding conductor 111 on itsback face.

As shown in FIG. 1C, the first U-shaped open transmission line resonator103 has bends so as to be in generally U shape, and is composed of threecontiguous connected transmission lines 11, 12 and 13. The transmissionlines 11 and 13 face each other in an almost parallel state, and thetransmission line 12 connects one end 11A of the transmission line 11and one end 13A of the transmission line 13. The transmission line 12 isbent at almost right angles at the end 11A of the transmission line 11and the end 13A of the transmission line 13.

Moreover, as shown in FIG. 1D, the crank-shaped open transmission lineresonator 104 has bends so as to be in generally crank shape, and iscomposed of three contiguous transmission lines 17, 18 and 19. Thetransmission lines 17 and 19 extend in an almost parallel state, and thetransmission line 18 connects one end 17A of the transmission line 17and one end 19B of the transmission line 19. The transmission line 18 isbent at almost right angles at the end 17A of the transmission line 17and the end 19B of the transmission line 19.

Moreover, as shown in FIG. 1E, the second U-shaped open transmissionline resonator 105 has bends so as to be in generally U shape, and iscomposed of three contiguous transmission lines 14, 15 and 16. Thetransmission lines 14 and 16 face each other in an almost parallelstate, and the transmission line 15 connects one end 14A of thetransmission line 14 and one end 16A of the transmission line 16. Thetransmission line 15 is bent at almost right angles at the end 14A ofthe transmission line 14 and the end 16A of the transmission line 16.

In this embodiment, each of the first U-shaped open transmission lineresonator 103, the second U-shaped open transmission line resonator 105and the crank-shaped open transmission line resonator 104 has a linelength which is approximately half an equivalent wavelength of apassband center frequency component.

Moreover, as shown in FIG. 1A, a section 101B of the first input/outputtransmission line 101 serving as the input line is in parallel proximityto the transmission line 11 of the first U-shaped open transmission lineresonator 103 with a specified gap therebetween and iselectromagnetically coupled to the transmission line 11. It is to benoted that the first input/output transmission line 101 serving as theinput line is composed of a section 101A and the section 101B, and thesection 101B extends from one end of the section 101A at almost rightangles to the section 101A.

The transmission line 13 of the first U-shaped open transmission lineresonator 103 and the transmission line 19 of the crank-shaped opentransmission line resonator 104 are arranged in parallel proximity toeach other with a specified gap therebetween so that portions of thesetransmission lines are electromagnetically coupled to each other.

Moreover, the transmission line 17 of the crank-shaped open transmissionline resonator 104 and the transmission line 14 of the second U-shapedopen transmission line resonator 105 are arranged in parallel proximityto each other with a specified gap therebetween so as to beelectromagnetically coupled to each other. Further, the transmissionline 16 of the second U-shaped open transmission line resonator 105 anda section 102B of the second input/output transmission line 102 servingas the output line are arranged in parallel proximity to each other witha specified gap therebetween so as to be electromagnetically coupled toeach other.

As shown in FIG. 1A, in the planar filter in the first embodiment, thecrank-shaped open transmission line resonator 104 is arranged in betweenthe first input/output transmission line 101 and the second input/outputtransmission line 102. Further, the transmission line 18 of thecrank-shaped open transmission line resonator 104 extends in almostparallel to the section 101A of the first input/output transmission line101 and a section 102A of the second input/output transmission line 102.In FIG. 1A, the direction along which the sections 101A, 102A extend isreferred to as “X direction”, and the direction perpendicular to the Xdirection is referred to as “Y direction”. Moreover, the transmissionlines 17 and 19 of the crank-shaped open transmission line resonator 104extend in opposite directions from both ends of the transmission line 18at almost right angles to the transmission line 18. Further, on bothsides of the transmission line 18 of the crank-shaped open transmissionline resonator 104 in the Y direction, the first U-shaped opentransmission line resonator 103 and the second U-shaped opentransmission line resonator 105 are disposed. The first U-shaped opentransmission line resonator 103 and the second U-shaped opentransmission line resonator 105 face each other in the Y direction, withtheir open ends displaced in the X direction.

Moreover, as shown in FIG. 1A, in the first embodiment, the gap betweenthe section 101B of the first input/output transmission line 101 and thetransmission line 11 of the first U-shaped open transmission lineresonator 103 is smaller than the gap between the transmission line 13of the first U-shaped open transmission line resonator 103 and thetransmission line 19 of the crank-shaped open transmission lineresonator 104. Further, the gap between the section 102B of the secondinput/output transmission line 102 and the transmission line 16 of thesecond U-shaped open transmission line resonator 105 is smaller than thegap between the transmission line 14 of the second U-shaped opentransmission line resonator 103 and the transmission line 17 of thecrank-shaped open transmission line resonator 104.

According to the thus-constructed planar filter, having the first andsecond U-shaped open transmission line resonators 103, 105 which arebent in U shape and the crank-shaped open transmission line resonator104 which is bent in crank shape makes it possible to decrease the areato be actually occupied by the filter on the dielectric substrate 110.This allows downsizing of a device having the planar filter.

Further, in the first embodiment, the first and second U-shaped opentransmission line resonators 103, 105 and the crank-shaped opentransmission line resonator 104 have a line length which is half anequivalent wavelength of a passband center frequency component, whichmakes it possible to enhance the wave filtration characteristic.

Further, in the first embodiment, the shape and arrangement of thethus-structured first and second U-shaped open transmission lineresonators 103, 105 and the crank-shaped open transmission lineresonator 104 make it possible to decrease the area on the dielectricsubstrate 110 to be actually occupied by the filter and enhance theattenuation characteristic, which in turn enables downsizing of a deviceemploying the planar filter.

In other words, according to the embodiment, the shapes and arrangementof the above-described resonators make it possible to realize a filterthat is equal to the conventional filter in terms of functions, andstill allows compact integration with an IC (Integrated Circuit).

Although in the embodiment, the first U-shaped open transmission lineresonator 103, the second U-shaped open transmission line resonator 105and the crank-shaped open transmission line resonator 104 areconstructed by angularly bending straight lines, the straight lines maybe gently bent in a curved shape, or corners of the bent straight linesmay be cut off.

Further, although in the embodiment, the transmission lines 11 to 13,the transmission lines 14 to 16 and the transmission lines 17 to 19 aremicro strip lines, they may be strip lines, suspended lines or coplanarlines. Moreover, although in the embodiment, the first input/outputtransmission line 101 serves as the input line and the secondinput/output transmission line 102 serves as the output line, the firstinput/output transmission line 101 may serve as the output line and thesecond input/output transmission line 102 may serve as the input line.

Second Embodiment

FIGS. 2A and 2B show a planar filter in the second embodiment of thepresent invention. FIG. 2A is a plan view, and FIG. 2B is a crosssectional view taken along line B-B′ in FIG. 2A.

The planar filter in the second embodiment has a first input/outputtransmission line 201 serving as an input line, a second input/outputtransmission line 202 serving as an output line, a first U-shaped opentransmission line resonator 203, a second U-shaped open transmissionline resonator 205 and a crank-shaped open transmission line resonator204, which are formed on a semi-insulative gallium arsenide substrate210 of a thickness of 70 μm. As shown in FIG. 2B, the semi-insulativegallium arsenide substrate 210 has a grounding conductor 211 on its backface.

As shown in FIG. 2C, the first U-shaped open transmission line resonator203 has bends so as to be in generally U shape, and is composed of threecontiguous connected transmission lines 21, 22 and 23. The transmissionlines 21 and 23 face each other in an almost parallel state, and thetransmission line 22 connects one end 21A of the transmission line 21and one end 23A of the transmission line 23. The transmission line 22 isbent at almost right angles at the end 21A of the transmission line 21and the end 23A of the transmission line 23.

Moreover, as shown in FIG. 2D, the crank-shaped open transmission lineresonator 204 has bends so as to be in generally crank shape, and iscomposed of three contiguous transmission lines 27, 28 and 29. Thetransmission lines 27 and 29 extend in an almost parallel state, and thetransmission line 28 connects one end 27A of the transmission line 27and one end 29B of the transmission line 29. The transmission line 28 isbent at almost right angles at the end 27A of the transmission line 27and the end 29B of the transmission line 29.

Moreover, as shown in FIG. 2E, the second U-shaped open transmissionline resonator 205 has bends so as to be in generally U shape, and iscomposed of three contiguous transmission lines 24, 25 and 26. Thetransmission lines 24 and 26 face each other in an almost parallelstate, and the transmission line 25 connects one end 24A of thetransmission line 24 and one end 26A of the transmission line 26. Thetransmission line 25 is bent at almost right angles at the end 24A ofthe transmission line 24 and the end 26A of the transmission line 26.

In the second embodiment, the transmission lines 21-29 each have athickness of 10 μm and a width of 30 μm. The transmission lines 21, 23,24 and 26 each have a center length of 385 μm, the transmission lines 22and 25 each have a center length of 180 μm, the transmission lines 27and 29 each have a center length of 275 μm, and the transmission line 28has a center length of 360 μm. Each of the first U-shaped opentransmission line resonator 203, the second U-shaped open transmissionline resonator 205 and the crank-shaped open transmission line resonator204 has a line length which is approximately half an equivalentwavelength of a passband center frequency component.

Moreover, as shown in FIG. 2A, a section 201B of the first input/outputtransmission line 201 serving as the input line is in parallel proximityto the transmission line 21 of the first U-shaped open transmission lineresonator 203 with a gap of 10 μm therebetween so as to beelectromagnetically coupled to the transmission line 21. It is to benoted that the first input/output transmission line 201 serving as theinput line is composed of a section 201A and the section 201B, and thesection 201B extends from one end of the section 201A at almost rightangles to the section 201A.

The transmission line 23 of the first U-shaped open transmission lineresonator 203 and the transmission line 29 of the crank-shaped opentransmission line resonator 204 are arranged in parallel proximity toeach other with a gap of 60 μm therebetween so that portions of thesetransmission lines are electromagnetically coupled to each other.

Moreover, the transmission line 27 of the crank-shaped open transmissionline resonator 204 and the transmission line 24 of the second U-shapedopen transmission line resonator 205 are arranged in parallel proximityto each other with a gap of 60 μm therebetween so as to beelectromagnetically coupled to each other. Further, the transmissionline 26 of the second U-shaped open transmission line resonator 205 anda section 202B of the second input/output transmission line 202 servingas the output line are arranged in parallel proximity to each other witha gap of 10 μm therebetween so as to be electromagnetically coupled toeach other.

As shown in FIG. 2A, in the planar filter in the second embodiment, thecrank-shaped open transmission line resonator 204 is arranged in betweenthe first input/output transmission line 201 serving as the input lineand the second input/output transmission line 202 serving as the outputline. Further, the transmission line 28 of the crank-shaped opentransmission line resonator 204 extends in almost parallel to thesection 201A of the first input/output transmission line 201 serving asthe input line and a section 202A of the second input/outputtransmission line 202 serving as the output line. In FIG. 2A, thedirection along which the sections 201A, 202A extend is referred to as“X direction”, and the direction perpendicular to the X direction isreferred to as “Y direction”. Moreover, the transmission lines 27 and 29of the crank-shaped open transmission line resonator 204 extend inopposite directions from both ends of the transmission line 28 at almostright angles to the transmission line 28. Further, on both sides of thetransmission line 28 of the crank-shaped open transmission lineresonator 204 in the Y direction, the first U-shaped open transmissionline resonator 203 and the second U-shaped open transmission lineresonator 205 are disposed. The first U-shaped open transmission lineresonator 203 and the second U-shaped open transmission line resonator205 face each other in the Y direction, without displacement of theiropen ends in the X direction.

The second embodiment is different from the first embodiment in that ina region V1 surrounded with a dotted line in FIG. 2A, a part 201A-1adjacent to the section 201B of the section 201A of the transmissionline 201 serving as the input line is arranged in parallel proximity toan end section 28A of the transmission line 28 of the crank-shaped opentransmission line resonator 204 with a gap of 60 μm therebetween so thatthey are electromagnetically coupled to each other. Moreover, in aregion V2 surrounded with a dotted line in FIG. 2A, a part 202A-1adjacent to the section 202B of the section 202A of the transmissionline 202 serving as the output line is arranged in parallel proximity toan end section 28B of the transmission line 28 in the crank-shaped opentransmission line resonator 204 with a gap of 60 μm therebetween so thatthose parts are electromagnetically coupled to each other.

According to the thus-constructed second embodiment, in addition to afirst signal transmission route on which a signal is transmitted in theorder of the transmission line 201 serving as the input line, the firstU-shaped open transmission line resonator 203, the crank-shaped opentransmission line resonator 204, the second U-shaped open transmissionline resonator 205 and the transmission line 202 serving as the outputline, there is formed a second signal transmission route on which asignal is transmitted in the order of the transmission line 201 servingas the input line, the crank-shaped open transmission line resonator 204and the transmission line 202 serving as the output line. This allowsmutual cancellation of signals in an attenuation band in the closevicinity of the pass band. Therefore, a large attenuation characteristiccan be obtained in a frequency band which requires attenuation.

FIG. 3 illustrates the transmission characteristic of the planar filterin the second embodiment by a transmission characteristic curve W1 drawnwith a solid line. A transmission characteristic curve W2 drawn with abroken line in FIG. 3 shows the transmission characteristic of theconventional planar filter. It is to be noted that the planar filter ofthe second embodiment and the conventional planar filter were formedthrough the same process with use of the same substrates. As is clearfrom the comparison between the transmission characteristic curve W1 andthe transmission characteristic curve W2, a passing loss within thepassband in the second embodiment is almost identical to that in theconventional example, but within the attenuation band in the range of 47to 57 GHz, a larger attenuation characteristic was obtained in thesecond embodiment than in the conventional example. In terms of thecharacteristic shown in FIG. 3, at a frequency of 50 GHz for example, anabsolute value of a transmission coefficient S21 (S parameter) in thesecond embodiment is larger by 5 (dB) than that in the conventionalexample, as indicated by reference symbol Y.

Thus, the planar filter in the second embodiment can achieve sufficientwave filtering performance while being more compact than theconventional planar filter.

Now, in order to demonstrate the effect of the electromagnetic couplingin the region V1 and the region V2 in FIG. 2A, FIG. 7 shows a passingcharacteristic of the filter in the case where the gap length in each ofthe region V1 and the region V2 is changed. In FIG. 7, transmissioncharacteristic Y2, which is equal to the transmission characteristic W1in FIG. 3, is a transmission characteristic of the planar filter inwhich the gap length in each of the region V1 and the region V2 is 60μm. Further, in FIG. 7, transmission characteristic Y3 is a transmissioncharacteristic of the planar filter in which the gap length in each ofthe region V1 and the region V2 is 30 μm. Further, transmissioncharacteristic Y4 is a transmission characteristic of the planar filterin which each of the gap length in the region V1 and the region V2 is 10μm.

The gap length was changed by fixing the positions of the open ends ofthe section 201B of the transmission line 201 and the section 202B ofthe transmission line 202 in FIG. 2A and by translating the section 201Aand the section 202A. Moreover, characteristic Y0 shown in FIG. 7 is atransmission characteristic in the case where the input/outputtransmission lines 201, 202 and the crank-shaped open transmission lineresonator 204 are intentionally arranged so as not to beelectromagnetically coupled to each other, namely, characteristic Y0 isa transmission characteristic of the planar filter having the FIG. 1construction described in connection with the first embodiment.

As shown in FIG. 7, with the decreasing gap length in the region V1 andthe region V2, the electromagnetic coupling between the input/outputtransmission lines 201, 202 and the crank-shaped open transmission lineresonator 204 is increased, and a larger attenuation pole is formed inthe frequency range of 51 to 54 GHz, whereas in the frequency range of51 GHz or lower, the attenuation characteristic is deteriorated. AS isapparent, optimizing the gap length in the region V1 and the region V2allows adjustment of the attenuation characteristic in a desiredfrequency band in conformity to a target specification.

Although in the second embodiment, the first U-shaped open transmissionline resonator 203, the second U-shaped open transmission line resonator205 and the crank-shaped open transmission line resonator 204 areconstructed by angularly bending straight lines, the straight lines maybe gently bent in a curved shape, or corners of the bent straight linesmay be cut off.

Further, although in the second embodiment a semi-insulating galliumarsenide substrate is used as the dielectric substrate, other substratesmade of semiconductor such as indium phosphorus, gallium nitride,silicon and so on may be employed. Further, the planar filter of thepresent invention can be constructed by employing a substrate made ofceramics such as alumina or glass, or a substrate made of a resin suchas Teflon (trade name of polytetrafluoroethylene made by DuPont).

Further, although in the second embodiment the transmission lines aremicro strip lines, they may be strip lines, suspended lines or coplanarlines. Moreover, although in the second embodiment, the firstinput/output transmission line 201 serves as the input line and thesecond input/output transmission line 202 serves as the output line, thefirst input/output transmission line 201 may be used as the output line,and the second input/output transmission line 202 as the input line.Also, although the second embodiment is an example of a millimeter waveband planar filter, the present invention is also applicable tomicrowave band planar filters.

Third Embodiment

Next, FIGS. 4A and 4B show a planar filter-integrated even-harmonicmixer device that is a semiconductor device as a third embodiment of thepresent invention. FIG. 4A is a plan view and FIG. 4B is a crosssectional view taken along line C-C′ in FIG. 4A. The planarfilter-integrated even-harmonic mixer device in the third embodiment isformed by integrating a planar filter 301 according to the secondembodiment shown in FIG. 2 with an even-harmonic mixer 300 on asemiconductor substrate.

The even-harmonic mixer device in the third embodiment is anup-converter even-harmonic mixer device for converting anintermediate-frequency signal to a high-frequency signal. The mixerdevice receives an intermediate-frequency signal (having a frequency(f_(IF))) and a local oscillation signal (having a frequency (f_(LO))),and mixes the intermediate-frequency signal and the local oscillationsignal to output a high-frequency signal (having a frequency (f_(RF))).The frequency (f_(IF)), the frequency (f_(LO)) and the frequency(f_(RF)) have a relationship expressed by the following equation (1):f _(RF)=2×f _(LO) +f _(IF).  (1)

In the third embodiment, it is assumed that the frequency f_(LO) of thelocal oscillation signal is 27.769 GHz, the frequency f_(IF) of theintermediate-frequency signal is 3.471 to 5.546 GHz, and that thefrequency f_(RF) of the high-frequency signal is 59.01 to 61.085 GHz.The gallium arsenide substrate has a size of approximately 1.5 mm×1.0mm, and the substrate has a thickness of 70 μm.

The planar filter-integrated even-harmonic mixer device of the thirdembodiment has the even-harmonic mixer 300, a phase adjustmenttransmission line 302 and the planar filter 301.

The even-harmonic mixer 300 is connected to between anintermediate-frequency signal terminal 309 and the phase adjustmenttransmission line 302. The even-harmonic mixer 300 has an MIM (MetalInsulator Metal) capacitor 305 connected to the intermediate-frequencysignal terminal 309, an intermediate-frequency signal transmission line304 connecting the MIM capacitor 305 to an open stub 30, and ananti-parallel diode pair 306 connected to the open stub 303. Further,the even-harmonic mixer 300 has a local oscillation signal transmissionline 308 connecting the anti-parallel diode pair 306 to a localoscillation signal terminal 311, and a short stub 307 connecting thelocal oscillation signal transmission line 308 to a pad 313. As shown inFIG. 4B, the pad 313 is connected via a through hole 312 formed througha gallium arsenide substrate 314 to a grounding conductor 315 formed onthe back face of the gallium arsenide substrate 314. The anti-paralleldiode pair 306 is formed on the gallium arsenide substrate 314 through asemiconductor process.

Moreover, each of the short stub 307 and the local oscillation signaltransmission line 308 has a line width of 50 μm so that thecharacteristic impedance becomes approximately 50Ω. Moreover, theintermediate frequency signal transmission line 304 is formed to have aline width of 20 μm so that the characteristic impedance becomesapproximately 70Ω. The stub 307, the transmission line 304 and thetransmission line 308 are properly bent to reduce the total size.

The length of the open stub 307 including the length of the through hole312 and the pad 313 is set so as to be about one quarter of thewavelength of the local oscillation signal of frequency f_(LO). The MIMcapacitor 305 is set to 0.4 pF so that the capacitor displays highimpedance with respect to the intermediate frequency signal (having afrequency of f_(IF)) and low impedance with respect to thehigh-frequency signal(having a frequency of f_(RF)).

Further, the phase adjustment transmission line 302 is almost equivalentto a transmission line of 50Ω, and has a function to delay only thephase without changing the amplitude. The phase adjustment transmissionline 302 is adjusted so that when an inputted signal is at the frequencyf_(LO), the impedance on the right-hand side as viewed from connectionpoint X in FIG. 4A (i.e., the side of the phase adjustment transmissionline 302 and the filter 301) becomes almost zero. Therefore, theconnection point X in the phase adjustment transmission line 302 can beregarded as equivalent to grounding with respect to the signal of thefrequency f_(LO).

Further, a local oscillation signal of the frequency f_(LO) inputtedfrom the local oscillation signal terminal 311 is supplied through thelocal oscillation signal transmission line 308 to the anti-paralleldiode pair 306. Because the short stub 307 has a length set to be aquarter wavelength with respect to the signal of the frequency f_(LO),the stub becomes equivalent to being open with respect to the signal ofthe frequency f_(LO), and this means that nothing is practicallyconnected.

Also, because the impedance on the right-hand side as viewed from theconnection point X in FIG. 4A is almost zero with respect to the signalof the frequency f_(LO), the connection point X practically almostsatisfies the condition of grounding with respect to the signal of thefrequency f_(LO). Therefore, all the voltage of the local oscillationsignal of the frequency f_(LO) inputted from the local oscillationsignal terminal 311 is applied to the anti-parallel diode pair 306.

The local oscillation signal inputted from the local oscillation signalterminal 311 and the intermediate-frequency signal of the frequencyf_(IF) inputted from the intermediate-frequency signal terminal 309 aremixed in the anti-parallel diode pair 306, as a result of which signalswith various frequency components are generated.

Among these signals with various frequency components, only a signalwith a frequency component which satisfies the equation (1), e.g.,(f_(RF)=2×f_(LO)+f_(IF)), passes through the bandpass filter 301.Unnecessary signals having other frequency components, which do notsatisfy the equation (1), cannot pass through the bandpass filter 301,but are reflected thereby. Moreover, among these unnecessary signals,signals particularly high in signal intensity, i.e., signal waves at afrequency of from 49.992 GHz to 52.067 GHz, or (2×f_(LO)+f_(IF)) can beconsiderably attenuated by the planar filter 301 having a characteristicW1 shown by solid line in FIG. 3.

As a result, in the planar filter-integrated even-harmonic mixer devicein the third embodiment, only the signal having the frequency of f_(RF)(=2×f_(LO)+f_(IF)) is outputted from a high-frequency signal terminal310. It is to be noted that the open stub 303 is intended to achievematching between the even-harmonic mixer 300 and the planar filter 301with respect to the signal of the frequency f_(RF).

Since the intermediate-frequency signal transmission line 304 has alength set to be a quarter wavelength of the signal of the frequencyf_(RF), the transmission line becomes equivalent to being open withrespect to the signal of the frequency f_(RF), and this means thatnothing is practically connected. Consequently, the signal of thefrequency f_(RF) is not outputted from the intermediate-frequency signalterminal 309.

Further, if the frequency f_(IF) of the intermediate-frequency signal ismuch smaller than the frequency f_(RF) of the high-frequency signal,then the following equation (2) is satisfied:f _(RF)≈2×f _(LO)  (2)

Therefore, the short stub 307 comes to have approximately halfwavelength with respect to the high-frequency signal of the frequencyf_(RF), and therefore becomes roughly equivalent to grounding for thehigh-frequency signal of the frequency f_(RF). Therefore, thehigh-frequency signal of the frequency f_(RF) is not outputted from thelocal oscillation signal terminal 311.

FIG. 8 shows one example of the characteristic of the even-harmonicmixer device. In FIG. 8, the horizontal axis represents a frequency ofthe IF signal, i.e., an intermediate-frequency signal f_(IF), whereasthe vertical axis represents a conversion gain. More particularly, thegraph shows a ratio of output power to input power in the IF signal. InFIG. 8, M1 represents conversion gains of the unnecessary waves havingfrequencies of (2×f_(LO)−f_(IF)), while M2 denotes conversion gains ofthe desired waves having the frequencies of (2×f_(LO)+f_(IF)).

Within the desired intermediate-frequency band of from 3.471 GHz to5.546 GHz, the conversion gain M2 is approx. −12 dB, whereas theconversion gain M1 is −45 dB or lower, and therefore the differencebetween the conversion gains is 33 dB or larger. This indicates thatoutput of the unnecessary wave is as small as 1/1000 of output of thedesired wave or less.

Thus, in the planar filter-integrated even-harmonic mixer device in thethird embodiment, integrating the planar filter 301 with theeven-harmonic mixer 300 on the same chip allows realization of asemiconductor device with extremely small output of the unnecessarywave. Moreover, since the power loss at the connection point X betweenthe even-harmonic mixer 300 and the planar filter 301 can be minimized,the performance is increased.

Further, as in the case of using the phase adjustment transmission line302 to implement equivalent grounding for the local oscillation signalof the frequency f_(LO), part of the characteristic of the planar filter301 in the present invention may be utilized in designing theeven-harmonic mixer 300, which makes it possible to simplify the circuitand realize a downsized semiconductor device.

It is to be noted that although the semi-insulative gallium arsenidesubstrate 314 is used as a semiconductor substrate in the embodiment,other semiconductor substrates made of indium phosphorus, galliumnitride, silicon and so on may be employed. Moreover, although theplanar filter is integrated with the even-harmonic mixer on thesemiconductor substrate in the embodiment, the planar filter may beintegrated with a fundamental wave mixer, and a circuit includingtransistors such as amplifiers may be also mounted on the same chip.

Further, although description of the mixer device has been given of thefunction as an up-converter for converting an intermediate-frequencysignal to a high-frequency signal in the embodiment, the mixer devicemay be used as a down-converter for converting a high-frequency signalto an intermediate-frequency signal.

Fourth Embodiment

Next, FIG. 5 shows a construction of a radio unit as a fourth embodimentof the present invention. The radio unit in the fourth embodiment is aradio relay device including a planar filter-integrated harmonic mixer506 according to the third embodiment.

The radio relay device in the fourth embodiment has an up-converter 501and a down-converter 521. The up-converter 501 up-converts a TVbroadcast signal in a UHF band to a signal in a millimeter wave band andsends the signal wirelessly, whereas the down-converter 521 (receiver)receives the signal and down-converts the signal to a signal in theoriginal UHF band.

The up-converter 501 has a bandpass filter 502 with a passband of from470 to 770 MHz, a bandpass filter 503 with a passband of from 3.941 to4.241 GHz, a bandpass filter 504 with a passband of 3.471 GHz and abandpass filter 505 with a passband of 27.769 GHz.

Further, the up-converter 501 also has a phase locked oscillator 507having an oscillation frequency of 3.471 GHz, an octupler 508, a mixer509, amplifiers 511, 512, 513, dividers 514, 515, a combiner 516, anattenuator 517, an antenna 518, and the planar filter-integratedeven-harmonic mixer 506 according to the third embodiment.

The down-converter 521 has amplifiers 522, 523, a millimeter wave filter524, a bandpass filter 525 with a passband of from 470 to 770 MHz, amixer 526 and an antenna 527.

Description will be given of the operation of the radio relay device inthe fourth embodiment below.

First, in the up-converter 501, a local oscillation signal of 3.471 GHzoutputted from the phase locked oscillator 507 is divided by the divider514 into two signals after passing through the bandpass filter 504, andone of the two signals is inputted into the divider 515, while the othersignal is inputted into the octupler 508. Next, in the divider 515, theinput signal is further divided into two signals, and one signal isinputted into the mixer 509, while the other signal is inputted into thecombiner 516 via the attenuator 517.

Moreover, the signal inputted into the octupler 508 is octupled tobecome a signal of 27.769 GHz, and after passing the bandpass filter505, the signal is inputted into a local oscillation signal terminal ofthe planar filter-integrated even-harmonic mixer 506.

Moreover, a UHF signal of a frequency of 470-770 MHz passes through thebandpass filter 502 and the amplifier 511, and then in the mixer 509,the signal is up-converted to a signal of 3.941-4.241 GHz by a localoscillation signal of 3.471 GHz. Then, after passing the bandpass filter503 and the amplifier 512, the signal is combined with a signal of 3.471GHz in the combiner 516.

As a result, the signal in the band of from 3.941 to 4.241 GHz and thesignal of 3.471 GHz are outputted from the combiner 516. These signalsare inputted into the intermediate-frequency signal terminal 309 of theplanar filter-integrated even-harmonic mixer 506, and mixed with a localoscillation signal of 27.769 GHz so as to be up-converted to a signal of59.01 GHz and a signal of 59.48-59.78 GHz. After an unnecessary signalis removed by the planar filter 301 in the planar filter-integratedeven-harmonic mixer 506, the remaining signal is amplified in theamplifier 513 and emitted to the air from the antenna 518 as amillimeter wave signal M.

In the down-converter 521, the signal in the band of 59.48-59.78 GHz andthe signal of 59.01 GHz are received by the antenna 527, and areinputted into the mixer 526 via the amplifier 522 and the millimeterwave filter 524. In the mixer 526, the signal in the signal band of from59.48 to 59.78 GHz and the signal of 59.01 GHz are mixed, and only asignal in a band of from 470 to 770 MHz is extracted by the bandpassfilter 525 and amplified by the amplifier 523.

As a result, a signal is reproduced, which has a frequency in a wavebandthat agrees with the waveband (470-770 MHz) of the signal inputted intothe up-converter 501.

According to the radio relay device of the fourth embodiment, inclusionof the planar filter-integrated even-harmonic mixer 506 of the presentinvention in the device allows reduction in component parts count of theup-converter 501 and downsizing of the device, as well as reduction inradiation of the unnecessary wave. It goes without saying thatindependent use of the planar filter 301 in the second embodimentinstead of use of the planar filter-integrated even-harmonic mixer 506still has a large effect on downsizing of the device and reduction inemission of the unnecessary wave.

Although description has been given of the radio relay device as oneexample of the radio unit in the fourth embodiment, the radio unit canbe implemented as a radio communication device.

Fifth Embodiment

Next, FIG. 9 shows the construction of a radio unit as a fifthembodiment of the present invention. The radio unit of the fifthembodiment is a radio relay device and includes a planar filter of thepresent invention.

The radio relay device has an up-converter 601 and a down-converter 621.The up-converter 601 up-converts a TV broadcast signal in a UHF band toa signal in a millimeter wave band and sends the signal wirelessly,whereas the down-converter 621 receives the signal and down-converts thesignal to a signal in the original UHF band.

The up-converter 601 has a bandpass filter 602 with a passband of from470 to 770 MHz, a bandpass filter 603 with a passband of from 3.941 to4.241 GHz and a bandpass filter 604 with a passband of 3.471 GHz.

Further, the up-converter 601 also has a phase locked oscillator 607having an oscillation frequency of 3.471 GHz, an oscillator 605 havingan oscillation frequency of 27.769 GHz, a mixer 609, amplifiers 611,612, 613, a divider 615, a combiner 616, an attenuator 617, an antenna618, and a planar filter-integrated even-harmonic mixer 606 according tothe third embodiment.

The down-converter 621 has a bandpass filter 622 with a passband of from470 to 770 MHz, a bandpass filter 623 with a passband of from 3.941 to4.241 GHz and a bandpass filter 624 with a passband of 3.471 GHz. Thedown-converter 621 also has an oscillator 625 having an oscillationfrequency of 27.769 GHz, a mixer 629, amplifiers 631, 632, 633, 634, adivider 636, an antenna 627, and a planar filter-integratedeven-harmonic mixer 626 according to the third embodiment.

The planar filter-integrated even-harmonic mixer 606 included in theup-converter 601 and the planar filter-integrated even-harmonic mixer626 included in the down-converter 621 have the same construction.

Description will be given of the operation of the radio relay device inthe fifth embodiment below.

First, in the up-converter 601, an oscillation signal of 3.471 GHzoutputted from the phase locked oscillator 607 is divided by the divider615 into two signals after passing through the bandpass filter 604, andone of the two signals is inputted in the mixer 609 as a localoscillation signal, while the other signal is inputted as a referencesignal into the combiner 616 via the attenuator 617.

Moreover, a sine wave with a frequency of 27.769 GHz is produced in theoscillator 605, and inputted into a local oscillation signal terminal ofthe planar filter-integrated even-harmonic mixer 606.

Moreover, a UHF signal of a frequency of 470-770 MHz passes through thebandpass filter 602 and the amplifier 611, and then in the mixer 609,the signal is up-converted to a signal of 3.941-4.241 GHz by a localoscillation signal of 3.471 GHz. Then, after passing the bandpass filter603 and the amplifier 612, the signal is combined with the referencesignal of 3.471 GHz in the combiner 616.

As a result, the signal in the wave band of 3.941-4.241 GHz and thereference signal of 3.471 GHz are outputted from the combiner 616. Thesesignals are inputted into the intermediate-frequency signal terminal 309of the planar filter-integrated even-harmonic mixer 606, and mixed witha local oscillation signal of 27.769 GHz so as to be up-converted to asignal of 59.01 GHz and a signal in a waveband of from 59.48 GHz to59.78 GHz. After unnecessary signal components are removed by the planarfilter 301 in the planar filter-integrated even-harmonic mixer 606, theremaining signal is amplified in the amplifier 613 and emitted as amillimeter wave signal MM to the air from the antenna 618.

In the down-converter 621, the signal of 59.01 GHz and the signal in theband of 59.48-59.78 GHz are received at the antenna 627, and amplifiedin the amplifier 633 before being inputted into the mixer device 626. Inthe mixer device 626, the sine wave of 27.769 GHz produced in theoscillator 625, the signal of 59.01 GHz and the signal in the band offrom 59.48 GHz to 59.78 GHz are mixed and down-converted to a signal ina band of 3.941-4.241 GHz and a reference signal of 3.471 GHz.

These signals are amplified by the amplifier 632 and divided by thedivider 636 into two, and one of the two signals is inputted into theband filter 624 where only the reference signal of 3.471 GHz isextracted. The extracted signal is amplified by the amplifier 634 andthen inputted into a local oscillation signal terminal of the mixer 629.The other signal from the divider 636 is inputted into the bandpassfilter 623 where a signal in the band of 3.941-4.241 GHz is extracted,and the extracted signal is inputted into a high-frequency terminal ofthe mixer 629. In the mixer 629, the signal of the waveband of3.941-4.241 GHz is mixed with the reference signal of 3.471 GHz inputtedat the local oscillation signal terminal to thereby be down-converted,and after being amplified by the amplifier 631, the signal is inputtedinto the bandpass filter 622 by which only the signal in the band of470-770 MHz is extracted.

In the radio relay device in the fifth embodiment, the reference signalof 3.471 GHz produced by the phase locked oscillator 607 of theup-converter 601 is up-converted by the planar filter-integratedeven-harmonic mixer 606 and is down-converted by the planarfilter-integrated even-harmonic mixer 626. Consequently, the referencesignal having a frequency of 3.471 GHz produced by the phase lockedoscillator 607 and then up-converted is back to a signal of a frequencyof 3.471 GHz, although the latter signal includes phase noise from theoscillators 605 and 625.

Similarly, the TV broadcast signal wave is up-converted anddown-converted by the planar filter-integrated even-harmonic mixers 606and 626. Consequently, the TV broadcast signal wave also becomes asignal including phase noise from the oscillators 605 and 625. However,in the mixer 629 in the down-converter 621, the TV broadcast signal withphase noise is mixed with the aforementioned down-converted referencesignal of 3.471 GHz so that the phase noise is cancelled. Eventually,therefore, from the bandpass filter 622 of the down-converter 621, thereis reproduced a UHF band signal having a frequency which agrees with thefrequency of the UHF band signal inputted into the bandpass filter 602of the up-converter 601.

Moreover, in the down-converter 621, an input signal is divided into asignal in a frequency band of 3.941-4.241 GHz and a reference signal of3.471 GHz by the divider 636 and the bandpass filters 623, 624, and withonly the reference signal of 3.471 GHz amplified by the amplifier 634,the mixer 629 is driven in a linear region. This reduces distortion ofthe signal outputted from the down-converter 621, resulting in anincreased communication distance.

The scheme adopted in the radio relay device in the fifth embodiment,which is particularly effective in terrestrial digital TV broadcastingusing orthogonal frequency division multiplexing (OFDM), is also able towirelessly relay satellite communication/broadcasting IF signals of afrequency of about 1 to 2 GHz.

Further, the arrangement for canceling the phase noise has beendescribed by way of example in the fifth embodiment, although thefilter-integrated even-harmonic mixer having the planar filter of thepresent invention can also be used as a mixer for a common heterodynetransmitter and receiver using a microwave band or a millimeter waveband.

Further, as described in connection with the fifth embodiment, use ofthe planar filter-integrated even-harmonic mixer having the planarfilter 301 of the present invention allows reduction in component partscount of the up-converter 601 and the down-converter 621 and downsizingof the device, as well as reduction in radiation of the unnecessarywave.

Further, the arrangement of the fifth embodiment allows the planarfilter-integrated even-harmonic mixers 606 and 626 and the oscillators605 and 625 to be component parts common to the up-converter 601 and thedown-converter 621. Moreover, the millimeter wave amplifiers 613 and 633may also be component parts common to the up-converter 601 and thedown-converter 621. Therefore, it becomes possible to reduce lines ofmillimeter-wave parts which are expensive at present. It goes withoutsaying that independent use of the planar filter of the presentinvention, without use of the planar filter-integrated even-harmonicmixer, still has a large effect on downsizing of the device andreduction in radiation of the unnecessary wave.

Embodiments of the invention being thus described, it will be obviousthat the same may be varied in many ways. Such variations are not to beregarded as a departure from the spirit and scope of the invention, andall such modifications as would be obvious to one skilled in the art areintended to be included within the scope of the following claims.

1. A planar filter, comprising: a first U-shaped open transmission lineresonator; a second U-shaped open transmission line resonator; and acrank-shaped open transmission line resonator having a middle portionand first and second end portions at opposing ends of the middleportion, the first end portion being bent at substantially a right anglewith respect to the middle portion to extend in a first direction fromthe middle portion and the second end portion being bent atsubstantially a right angle with respect to the middle portion to extendin a second direction from the middle portion, opposed to the firstdirection.
 2. The planar filter as defined in claim 1, wherein the firstand second U-shaped open transmission line resonators and thecrank-shaped open transmission line resonator have a line length that ishalf an equivalent wavelength of a passband center frequency component.3. The planar filter as defined in claim 1, wherein the first and secondU-shaped open transmission line resonators and the crank-shaped opentransmission line resonator are arranged so as to be electromagneticallycoupled in an order of the first U-shaped open transmission lineresonator, the crank-shaped open transmission line resonator, and thesecond U-shaped open transmission line resonator.
 4. The planar filteras defined in claim 3, further comprising: a first input/outputtransmission line and a second input/output transmission line, whereinthe first input/output transmission line is arranged so as to beelectromagnetically coupled to the first U-shaped open transmission lineresonator, and the second input/output transmission line is arranged soas to be electromagnetically coupled to the second U-shaped opentransmission line resonator.
 5. The planar filter as defined in claim 4,wherein the first and second input/output transmission lines and thecrank-shaped open transmission line resonator are arranged such that apart of at least one of the first and second input/output transmissionlines and a part of the crank-shaped open transmission line resonatorare electromagnetically coupled to each other.
 6. The planar filter asdefined in claim 1, wherein the first and second U-shaped opentransmission line resonators and the crank-shaped open transmission lineresonator are formed on a semiconductor substrate.
 7. A semiconductordevice comprising the planar filter as defined in claim 1, wherein theplanar filter is integrated with a mixer on a semiconductor substrate.8. A radio unit comprising the planar filter as defined in claim 1.